Output stage and method of enhancing output gain

ABSTRACT

The present invention provides for an output stage which couples with an input stage and is configured to limit a reflection current which is reflected back into the input stage to enhanced an output voltage to drive a variety of loads. The present output stage limits the reflection current by compensating for at least one bias current at the input stage output. The output stage further reduced a quiescent current needed to maintain the output stage in an active state without adversely affecting the output voltage supplied to the load. The output stage includes a first and second current driving stage, a first and second current compensation circuit and an output circuit. The first and second current driving stages couple with both the input stage and the output circuit. The first and second current driving stages are configured to generate a first and second driving current, respectively, to drive the output circuit, wherein the first driving current is proportional to a first bias current and the second driving current is proportional to a second bias current. To limit reflection current the first current compensation circuit is configured to compensate for the first bias current while the second current compensation circuit is configured to compensate for the second bias current.

FIELD

[0001] This invention pertains to an output stage, and more particularly to an output stage and method for preventing offset voltage error and improving output stability and performance.

BACKGROUND

[0002] It is known in the prior art to provide some degree of output stage control for amplifier outputs. However, prior art amplifier output stages have several limiting features which affect the amplification of the amplifier input signal and fail to provide adequate control of the amplifier output.

[0003]FIG. 1 depicts an example of a prior art amplifier 120 with an output stage 122. One of the problems with prior art amplifier output stages is that they cause an excessively large reflection current (ΔI) to be reflected back into amplifier 120. The current reflection ΔI will be reflected back through transconductance amplifier 120 and produce a systematic input offset error voltage (V_(OS)) which can have significant effects on the performance of amplifier 120. The resulting offset error voltage V_(OS) is equivalent to the reflection current ΔI divided by the amplifier transconductance (g_(m)), V_(OS)=ΔI/g_(m). Therefore, any increase in ΔI will result in an unwanted increase in offset error voltage V_(OS).

[0004] In the prior art amplifier output stage 122, current I_(o) provides the driving current to the base of a source NPN transistor Q3 which in turn controls the current flow through transistor Q3 and thus output voltage V_(out). Because the base current I_(B) of a transistor provides control for the collector current I_(C) through the known beta factor relationship of I_(C)=I_(B)*β, I_(B3) provides control for the collector current I_(C3) of transistor Q3 and thus the output voltage V_(out). Therefore, to achieve a sufficiently large output voltage, V_(out), through transistor Q3, base current I_(B3) of transistor Q3 must be sufficiently large to drive I_(C3). To achieve a sufficiently large I_(B3), the base current of transistor Q1 (I_(B1)) must be sufficiently large (again, I_(C)=I_(B)*β) to generate a sufficiently large I_(o) current to drive transistor Q3.

[0005] This large base current I_(B1) causes a significant increase in the current at node N1. This increase at node N1 results in an increase in reflection current ΔI. The resulting increased ΔI is reflected back through transconductance amplifier 120 resulting in an increased offset error voltage V_(OS) which in turn affects the overall amplifier output voltage V_(out), depending on the closed-loop gain.

[0006] A similar analysis can be made with regard to driving transistor Q2 which drives sink output transistor Q4. To achieve a sufficiently large output current through transistor Q4, a base current (I_(B4)) of transistor Q4 must be sufficiently large to produce a sufficiently large collector current (I_(C4)) for transistor Q4. Therefore, I_(o) through driving transistor Q2 must be sufficient to drive sink output transistor Q4. This requires an increase in base current (I_(B2)) for transistor Q2 which in turn affects the current at node N1.

[0007] The prior art output stage 122 further affects the overall amplifier output voltage V_(out) because of the needed implementation of Q1 as a fast PNP transistor. To achieve a desired bandwidth and efficiency, Q1 needs to be a fast transistor. Therefore, the Q1 transistor is implemented with a vertical PNP configuration to improve response time. However, fast, vertical PNP transistors have a reduced β, which in turn reduces the driving current I_(o). Thus, to maintain I_(o) at a sufficiently large level to drive source transistor Q3, I_(B1) must be further increased. The further increase to I_(B1) causes an increase in the current at node N1, resulting in an increase in reflection current ΔI and thus an increase in offset error voltage V_(OS).

[0008] What is needed is an amplifier output stage which provides a sufficiently large driving current to the base of a source transistor without adversely increasing the reflection current ΔI. What is further needed is an output stage which will significantly reduce or eliminate any reflection current ΔI, where ΔI is due to a mismatch of beta factors of NPN transistors (β_(npn)) and PNP transistors (β_(pnp)).

SUMMARY

[0009] The present invention provides for an output stage which couples with an input stage and is configured to limit a reflection current which is reflected back through the input stage producing an offset error input voltage affecting the performance of the input stage. The present output stage limits the reflection current by compensating for at least one bias current at the input stage output. The output stage further reduces a quiescent current needed to maintain the output stage in an active state without adversely affecting the output voltage supplied to the load. The output stage includes a first and second current driving stage, a first and second current compensation circuit and an output circuit. The first and second current driving stages couple with both the input stage and the output circuit. The first current driving stage is configured to generate a first driving current to drive the output circuit, wherein the first driving current is proportional to a first bias current based on an output from the input stage. The second driving stage is configured to generate a second driving current to also drive the output circuit, wherein the second driving current is proportional to a second bias current based on the output of the input stage. To limit reflection current back into the input stage, the first current compensation circuit couples with the first driving stage and the input stage, and is configured to compensate for the first bias current. To further limit the reflection current back into the input stage, the second current compensation circuit couples with the second driving stage and the input stage, and is configured to compensate for the second bias current.

[0010] In one embodiment, the first current driving stage further includes a first current multiplier coupled with the output circuit, and is configured to drive the output circuit at a sufficient level while limiting the first bias current. The second current driving stage further includes a second current multiplier coupled with the output circuit, and is configured to drive the output circuit at a defined level while limiting the second bias current. The first current compensation circuit includes a first current mirror coupled with the first current driving stage to compensate for the first bias current and the second current compensation circuit includes a second current mirror coupled with the second current driving stage to compensate for the first bias current. The output stage further includes a clamping stage coupled with the input stage and the output circuit, and a feedforward path coupled between the input stage and the output circuit. The clamping stage is configured to maintain the output level of the output stage to be approximately equal to the input stage output, while the feedforward path is configured to stabilize the output stage output at high frequencies.

[0011] In one embodiment, the first current compensation circuit further includes a PNP sink transistor coupled with the first current driving stage to sink a first driving stage total collector current from the first current driving stage. The PNP sink transistor is configured to define a first compensation current to compensate for the first bias current wherein the first bias current is about equal to a total collector current of the first current driving stage divided by a PNP beta factor. Further, the second current compensation circuit includes an NPN source transistor coupled with the second current driving stage to supply a second driving stage total collector current from the second current driving stage. The NPN source transistor is configured to define a second compensation current to compensate for the second bias current wherein the second bias current is about equal to a total collector current of the second current driving stage divided by an NPN beta factor.

[0012] In accordance with the teachings of this invention a novel method and structure is taught which provides integration, access, and transportation software data and information networks capable of delivering adaptable, expandable, high volume, high performance, and fault tolerant capabilities.

BRIEF DESCRIPTION OF THE DRAWINGS

[0013] The invention, together with further advantages thereof, may best be understood by reference to the following description taken in conjunction with the accompanying drawings in which:

[0014]FIG. 1 depicts a prior art amplifier and amplifier output stage;

[0015]FIG. 2 is a block diagram of one embodiment of an output stage of the present invention;

[0016]FIG. 3 is a schematic diagram of one embodiment of an output stage;

[0017]FIG. 4 depicts one embodiment of an output circuit which includes a plurality of source transistors and a plurality of sink transistors;

[0018]FIG. 5 is a schematic diagram of one embodiment of an output stage including a clamping stage;

[0019]FIG. 6 depicts a simplified schematic diagram of one embodiment of an output stage of FIGS. 2 and 3, including the equivalent series RC feedforward path; and

[0020]FIG. 7 depicts a simplified schematic diagram of one embodiment of the feedforward path of FIG. 6 implemented through an RC network.

DETAILED DESCRIPTION

[0021]FIG. 2 depicts a block diagram of one embodiment of the present novel output stage 150. Output stage 150 provides for an output voltage V_(out) while substantially eliminating reflection current ΔI from reflecting back into substantially any input stage circuit where reflection current has an adverse effect. In the embodiment shown in FIG. 2, output stage 150 substantially eliminates reflection current ΔI from reflecting back into an amplifier input stage 120 substantially eliminating a systematic input offset voltage which is caused by the reflection current. The reduction or elimination of reflection current ΔI is achieved by offsetting base currents for first and second current driving stages 152 a and 152 b, and further by reducing the amount or size of driving current I_(o). As shown in FIG. 2, amplifier input stage 120 is coupled to output stage 150. Output stage 150 includes first and second driving stages 152 a-b coupled with amplifier 120 at node N1. First and second driving stages 152 a-b couple with and drive an output circuit 154 of output stage 150. First and second driving stages 152 a-b reduce the current at node N1, and thus the reflection current ΔI which reflects back into amplifier input stage 120, by reducing the size of the driving current I_(o), as is described in more detail below. Refection current ΔI is further reduced by compensating for bias currents applied to or from driving stages 152 a-b through current compensation circuits 160 a-b, as is described more fully below.

[0022] In one embodiment, to improve bandwidth and speed of the system, first driving circuit 152 a is at least partially implemented utilizing vertical PNP transistors while the second driving circuit 152 b is at least partially implemented utilizing NPN transistors. Thus, in one embodiment, the beta factor β_(pnp) for first driving circuit 152 a will not equal the beta factor β_(npn) for the second driving circuit. This mismatch of beta factors results in a mismatch in bias currents I_(B1) and I_(B2) resulting in a reflection current ΔI. To compensate for this mismatch, reflection current ΔI is further reduced by compensating for bias currents I_(B1) and I_(B2) into first and second driving stages 152 a-b, respectively, through a first and second current compensation circuit 160 a-b. First compensation circuit 160 a couples with first driving circuit and node N1 to compensate for bias current I_(B1) and second compensation circuit 160 b couples with second driving circuit and node N1 to compensate for bias current I_(B2). First current driving stage 152 a and second compensation circuit 160 b are coupled with a first voltage reference VCC, and second current driving stage 152 b and first compensation circuit 160 a further couple with a second voltage reference VDD.

[0023]FIG. 3 depicts a simplified schematic diagram of amplifier input stage 120 coupled with one embodiment of an output stage 150 of the present invention. In one embodiment, the present invention reduces, and preferable eliminates, the offset error voltage V_(OS) by compensating for the reflection current ΔI at the node N1. The reflection current ΔI is equal to the sum of the currents at node N1. Thus, reflection current Δis the sum of the bias or base currents I _(B21) and I_(B23) of both the first and second driving stages 152 a and 152 b, respectively. Reflection current Δcan be written as:

ΔI=I _(B21) −I _(B23),  (eq. 1)

[0024] where base current I_(B21) is the sum of the base currents I_(B21a) and I_(B21b) from transistors Q21 a and Q21 b, respectively, and base current I_(B23) is the sum of the base currents I_(B23a) and I_(B23b) of transistors Q23 a and Q23 b, respectively. Therefore, to eliminate reflection current, one can design first and second driving stages to have equivalent base currents. Alternatively, reflection current ΔI can be eliminated by compensating for each base current individually.

[0025] Because the base current of a transistor is substantially equal to the collector current divided by the beta factor β for that transistor, base current can be written as:

I _(B) =I _(C)/β.  (eq. 2)

[0026] Utilizing equation 2, equation 1 can be rewritten as:

ΔI=(I _(C21)/β_(pnp))−(I _(C23)/β_(npn))  (eq. 3)

[0027] As described above, because the transistors Q21A and Q21B of one embodiment of first driving stage 152 a are configured as fast, vertical PNP transistors to achieve the desired bandwidth and speed, the value of the beta factor β_(pnp) of these PNP transistors will generally not match the β_(npn) value of the NPN transistors Q23A and Q23B of second driving stage 152 b. Therefore, base current I_(B21) of first driving stage 152 a will generally not equal base current I_(B23) of second driving stage 152 b, resulting in a current at node N1 producing reflection current ΔI=I_(B21)−I_(B23).

[0028] In one embodiment, to compensate for the mismatch in base currents I_(B21) and I_(B23), output stage 150 includes first and second current compensation circuits 160 a and 160 b. Each current compensation circuit includes a current mirror circuit 162 a and 162 b. First current mirror 162 a further couples with the base of current sink transistor Q28 and receives the base current I_(B28) from sink transistor Q28. Sink transistor Q28 is configured to receive the driving stage total collector current I_(C21) from both transistors Q21 a and Q21 b of first driving stage 152 a. Second current mirror 162 b further couples with the base of current source transistor Q32 and supplies the base current to source transistor Q32. Source transistor Q32 is configured to supply the driving stage collector current I_(C23) for both transistors Q23 a and Q23 b of second driving stage 152 b.

[0029] Current mirrors 162 a-b of output stage 150 are configured in any convenient manner, for example a pair of base coupled transistors, or two pairs of base coupled transistors in a stacked configuration, as is known in the art. Current mirrors 162 a and 162 b depicted in FIG. 3 are simple representations of current mirrors, however, either one or both can be configured in alternative configurations. First current mirror 162 a includes a current control leg 164 and a current mirror output leg 166. In the embodiment depicted in FIG. 3, current control leg 164 includes current control transistor Q27 a and current mirror output leg 166 includes current mirror output transistor Q27 b. The base of control transistor Q27 a and the base of mirror output transistor Q27 b are both coupled with the collector of control transistor Q27 a. Thus each have substantially the same base current (I_(B27a)=I_(B27b)) or base to emitter voltage drop V_(BE). Because each transistor, Q27 a and Q27 b, has the same base current, each will have substantially the same collector current, I_(C27a)=I_(C27b), provided that each transistor has substantially the same β_(npn), which should be the case because both will be formed in substantially the same manner. Therefore, the current I_(C27b) in current mirror output leg 166 will mirror of the current I_(C27a) in control leg 164.

[0030] Second current mirror circuit 162 b follows the same logic as described for first current mirror circuit 162 a. Therefore, the current in current mirror output leg 172 mirrors the current in control leg 170.

[0031] Through current compensation circuits 160 a and 160 b, the mismatch between I_(B21) and I_(B23) is eliminated by providing a compensation current for each, thus reflection current ΔI is eliminated. First mirror circuit 162 a is configured to compensates for base current I_(B21) of transistors Q21 a and Q21 b, and second mirror circuit 162 b is configured to compensate of base current I_(B23) for transistors Q23 a and Q23 b. This is achieved by matching the first current mirror output I_(C27b) of first mirror circuit 162 a with base current I_(B21) of transistors Q21 a and Q21 b(I_(B21)=I_(B21a)+I_(B21b)), and matching the second current mirror output I_(C31b) of second mirror circuit 162 b with base current of transistors Q23 a and Q23 a(I_(B23)=I_(B23a)+I_(B23b)).

[0032] Because source transistor Q32 supplies the collector current I_(C23) to transistors Q23 a and Q23 b, I_(C23) is substantially equal to collector current I_(C32) of source transistor Q32. Thus, utilizing equation 2, I_(B23)=I_(C23)/β_(npn), where I_(C23)=I_(C23a)+I_(C23b), base current I_(B23) is substantially equal to:

I _(B23) =I _(C32)/β_(npn).  (eq. 4)

[0033] Again relying on equation 2, base current I_(B32) of source transistor Q32 is substantially equal to I_(C32)/β_(npn). Base current I_(B32) of source transistor Q32 is supplied by, and substantially equal to, collector current I_(C31a) of current control transistor Q31 a. Because collector current I_(C31b) of current mirror output transistor Q31 b equals collector current I_(C31a) of control transistor Q31 a, control mirror output current I_(C31b) equals:

I _(C31b) =I _(C32)/β_(npn).  (eq. 5)

[0034] Combining equations 4 and 5 (I_(B23)=I_(C32)/β_(npn), and I_(B31b)=I_(C32)/β_(npn)) results in:

I _(B23) =I _(C31b).  (eq.6)

[0035] Thus, second current compensation circuit 160 b compensates for base current I_(B23) of transistors Q23 a and Q23 b, and base current I_(B32) of source transistor Q32 defines a first compensation current.

[0036] Following similar calculations as described for the compensation of bias current I_(B23), bias current I_(B21) is compensated by first current compensation circuit 160 a. Utilizing equation 2, I_(B21)=I_(C21)/β_(pnp). Because sink transistor Q28 receives the total collector current I_(C21) from transistors Q21 a and Q21 b, total collector current I_(C21) is substantially equal to collector current I_(C28) of sink transistor Q28. Thus, base current I_(B21) is substantially equal to:

I _(B21) =I _(C28)/β_(pnp).  (eq. 7)

[0037] Again relying on equation 2, base current I_(B28) of sink transistor Q28 is substantially equal to I_(C28)/β_(pnp). Base current I_(B28) supplies, and is substantially equal to, collector current I_(C27a) of current control transistor Q27 a. Because collector current I_(C27a) of current mirror output transistor Q27 b mirrors collector current I_(C27a) of control transistor Q27 a, control mirror output current I_(C27b) is substantially equal to the collector current I_(C27a) of current control transistor Q27 b, therefore:

I _(C27a) =I _(C28)/β_(pnp).  (eq. 8)

[0038] Combining equations 7 and 8 results in:

I _(B21) =I _(C27a).  (eq. 9)

[0039] Thus, first current compensation circuit 160 a compensates for base current I_(B21) of transistors Q21 a and Q21 b, and base current I_(B28) of sink transistor Q28 defines a second compensation current.

[0040] As such, first and second current compensation circuits 162 a and 162 b provide further current, I_(C27b) and I_(C31b), to node N1, thus equation 1 can be rewritten to as:

ΔI=(I _(B21) −I _(C27b))−(I _(B23) −I _(C31b)),  (eq. 10)

[0041] and since I_(B21)=I_(C27a) and I_(B23)=I_(C31b), equation 10 results in ΔI=0. Therefore, resulting reflection current ΔI equals zero, and there is substantially no reflection current back into amplifier 120, thus eliminating offset error voltage V_(OS).

[0042] The present output stage 150 further provides for temperature and processing compensation to ensure continuous and accurate reflection current ΔI reduction and/or elimination. As is known in the art, changes in temperature and changes in circuit processing can alter or affect the operation of transistors. As such, changes in temperature or processing can alter the base currents I_(B21) and I_(B23) due to the changes in β and the operation of transistors Q21 and Q23 which may result in an altered or changed reflection current ΔI. However, because of the design of the current compensation circuits 154 a-b, changes in temperature or processing will also affect the transistors of current compensation circuits 154 a-b. Thus, current compensation circuits of the present invention maintain accurate compensation bias currents regardless of variations in temperature and processing. Thus providing a superior and more stable amplifier output.

[0043] In one embodiment, the present invention provides output circuit 154 with a sufficient driving current to generate a stable and accurate output voltage V_(out) while maintaining a quiescent current of first and second driving stages 152 a-b at a reduced level. As was described with respect to the prior art output stages, a sufficiently large current is needed to drive the output transistors to obtain a sufficient output current. However, in prior art designs, large bias currents are needed to generate the sufficient driving current to drive the output transistors. This can result in greater variations between bias currents and thus a larger reflection current ΔI along with increased power consumption.

[0044] In one embodiment the present invention reduces the quiescent current or idle current needed to maintain transistors Q21 and Q23 of current driving stages 152 a-b in an active state to prevent delays in forwarding the amplifier input stage output to the output circuit 154 without adversely affecting the level of the driving current at output circuit 154. Further, by reducing the idle current, the bias currents I_(B21) and I_(B23) are reduced thus reducing the level of compensation current needed to be generated by current compensation circuits 160 a-b. This reduced quiescent current and reduced compensation currents I_(C27b) and I_(C31b) results in a reduction in the total power consumption of output stage 150.

[0045] Still referring to FIG. 3, in one embodiment, output stage 150 reduces the quiescent current needed for first and second driving stages 152 a-b. Quiescent current or idle current is reduced by reducing the size of driving current I_(o1). Output circuit 154 of output stage 150 must be driven by a sufficiently large driving current to generate an adequate output voltage V_(out). However, a large driving current can adversely affect the reflection current back into amplifier 120 and requires a larger idle current to maintain transistors Q21 and Q23 in an active state to prevent delays as described above. Thus, in one embodiment, the driving current I_(o1) is limited to reduce the total quiescent current. In one embodiment, this is achieved by multiplying driving current I_(o1) through a current multiplier transistor Q22. Driving current I_(o1) passes through a first driving transistor Q20. Driving current I_(o1) drives the base current I_(B22) of current multiplier transistor Q22 depending upon the magnitude of the output of amplifier input stage, thus multiplying driving current I_(o1) to produce a first driving current I_(o2) to feed output source NPN transistor Q29 at a sufficient level resulting in a sufficient output voltage V_(out). Similarly, in the second driving stage 152 b, driving current I_(o1) from collector current of transistor Q24, is multiplied by multiplier transistor Q26 resulting in second driving current I_(o3) to drive the base of output sink PNP transistor Q30 resulting in a sufficient output voltage V_(out). By reducing the size of the driving current I_(o1), the bias or base currents of transistors Q21 a-b and Q23 a-b are significantly reduced. Thus the idle current is reduced and the net current at node N1 is reduced. The reduced net current at node N1 results in a reduced reflection current ΔI. Thus, the output stage 150 of the present invention reduces the idle current, the levels of compensation current needed and the overall power consumption by reducing the driving current I_(o1). A similar analysis can be provided for second driving transistor Q24 and second current multiplier transistor Q26 resulting in reduced idle current.

[0046] Further, the drive current I_(o1) is reduced in one embodiment without causing any further delay in the path between amplifier input stage 120 at node N1 to output circuit 154. The present invention includes only a single transistor delay between node N1 and output circuit 154. Thus, the present invention reduces driving current I_(o1) and thus the quiescent currents without any additional delay in the propagation of the output of amplifier input stage 120 at node N1 to output circuit 154.

[0047] Referring to FIG. 4, output circuit 154 of the present invention is configured in any convenient manner to provide amplifier output stage 150 output voltage V_(out). In one embodiment, output circuit 154 includes a plurality of output source transistors Q29 and a plurality of output sink transistors Q30. FIG. 4 depicts one embodiment of output circuit 154 which includes source transistors Q29 a-j, each coupled at the collector with a corresponding source resistance R20 a-j, and sink transistors Q30 a-j, each coupled at the collector with a corresponding sink resistance R21 a-j generating output voltage V_(out).

[0048]FIG. 5 depicts a simplified schematic diagram of one implementation of one embodiment of the novel output stage 150 of the present invention. Output stage 150 includes a clamping stage 190 to prevent saturation of transistors of current driving stages 152 a-b and other transistors associated with current driving stage 152 a-b. Preventing current driving stage transistor saturation further prevents signal delays in propagating the output of amplifier input stage 120 to output circuit 154 caused by the time needed for the transistors to become unsaturated. Clamping stage 190 also provides further control of output stage 150 and enhances the gain of output stage 150. Clamping stage 190 prevents transistor saturation if the voltage at node N1 rapidly transitions high or transitions low by maintaining the voltage level of node N1 to a voltage which is proportional to output voltage V_(out) In one embodiment, the voltage at node N1 is maintained to within one transistor base-emitter junction voltage drop V_(BE).

[0049] In one embodiment, clamping stage 190 includes an NPN clamping source transistor Q33 and a PNP clamping sink transistor Q34. The collector of clamping source transistor Q33 couples with power source VCC and the emitter couples with node N1. The base of transistor Q33 couples with the emitter of output sink transistor Q30 at node N2 of output circuit 154. The emitter of clamping sink transistor Q34 couples with node N1 and the collector couples with VDD. The base of clamping sink transistor Q34 couples with the collector of output source transistor Q29 at node N3 of output circuit 154. Clamping stage 190 is activated when the voltage at node N1 varies from the voltage at either node N2 or node N3 by more than a predefined voltage, for example, by more than the base-emitter voltage drop V_(BE) of transistor Q33 or Q34. This provides clamping of the output voltage of amplifier input stage 120 at node N1 with respect to the output voltage V_(out). If the voltage at node N1 rapidly falls resulting in a one V_(BE) drop at node N1 below the voltage at node N2, clamping source transistor Q33 will be activated maintaining the voltage at node N1 to within one V_(BE) of output voltage V_(out). If the voltage at node N1 rapidly rises to exceed the node voltage at node N3 by one V_(BE), clamping sink transistor Q34 is activated limiting the voltage at N1 again to within one V_(BE) of output voltage V_(out).

[0050] In one embodiment, the desired signal gain of output stage 150 is approximately equal to unity. However, as the frequency of the output signal from amplifier input stage 120 at node N1 increases the gain of output stage 150 may fall below unity. As the gain falls, capacitive loading effects will be reflected back through output stage 150 causing instability in the signal applied to the load and reducing the overall efficiency of output stage 150. To compensate for this instability, in one embodiment, output stage 150 of the present invention includes a feedforward path which improves the stability of the amplifier gain at higher frequencies. FIG. 6 depicts a simplified schematic diagram of the output stage 150, including the equivalent series RC feedforward path 210, coupled to amplifier 120 and a capacitive loading C_(L). Feedforward path 210 includes an equivalent feedforward resistance R_(FF) and an equivalent feedforward capacitance C_(FF) coupled between node N1 and the output voltage V_(out) of output stage 150. At higher frequencies, the effects of feedforward capacitance C_(FF) are reduced thus maintaining the gain of output stage 150 at approximately unity and improving high frequency performance and stability of output stage 150.

[0051] In one embodiment, feedforward path 210 is implemented through an RC network depicted in FIG. 7. As shown, feedforward path 210 includes first feedforward resistor R_(f1) which couples at one end to node N1 and at the other to second and third feedforward resistors R_(f2) and R_(f3). Second feedforward resistor R_(f2) further couples with first feedforward capacitance C_(f1). Third feedforward resistor R_(f3) further couples with second and third feedforward capacitance C_(f2) and C_(f3) coupled in parallel. First, second and third feedforward capacitance C_(f1), C_(f2), C_(f3), each couple with the output stage output voltage V_(out). Thus, providing a bypass for the amplifier output at high frequencies or frequencies above a predefined limit and providing stability to the amplifier gain.

[0052] As taught by the foregoing description and examples, an output stage for substantially eliminating reflection current ΔI is provided by the present invention. The foregoing description of specific embodiments and examples of the invention have been presented for the purpose of illustration and description, and although the invention has been illustrated by certain of the preceding examples, it is not to be construed as being limited thereby. They are not intended to be exhaustive or to limit the invention to the precise forms disclosed, and obviously many modifications, embodiments, and variations are possible in light of the above teaching. It is intended that the scope of the invention encompass the generic area as herein disclosed, and by the claims appended hereto and their equivalents. The invention was described in relation to substantially eliminating reflection current into an amplifier, however, the output stage of the present invention is equally applicable to substantially any circuit adversely affected by reflection current.

[0053] Having disclosed exemplary embodiments and the best mode, modifications and variations may be made to the disclosed embodiments while remaining within the scope of the present invention as defined by the following claims. 

What is claimed is:
 1. An output stage configured to receive an input stage output and to generate a stable output to a variety of loads, the output stage comprising: a first and second current driving stage, a first and second current compensation circuit and an output circuit; the first driving stage coupled with an input stage and the output circuit, and configured to generate a first driving current proportional with a first bias current based on the input stage output, to drive the output circuit; the second driving stage coupled with the input stage and the output circuit, and configured to generate a second driving current proportional with a second bias current based on the input stage output, to drive the output circuit; the first current compensation circuit coupled with the first driving stage and the input stage, and configured to compensate for the first bias current to limit a reflection current back into the input stage; and the second current compensation circuit coupled with the second driving stage and the input stage, and configured to compensate for the second bias current to further limit the reflection current back into the input stage.
 2. The output stage as claimed in claim 1, wherein: the first current driving stage includes a first current multiplier coupled with the output circuit, and configured to drive the output circuit at a predefined level while limiting the first bias current; and the second current driving stage including a second current multiplier coupled with the output circuit, and configured to drive the output circuit at a predefined level while limiting the second bias current.
 3. The output stage as claimed in claim 2, wherein: the first and second current multipliers include at least a first and second current multiplier transistor, respectively, each transistor having a base, collector and emitter; the emitter of the first current multiplier transistor coupled with the output circuit to supply a first drive current to drive the output circuit; and the emitter of the second current multiplied transistor coupled with the output circuit to draw a second drive current to drive the output circuit.
 4. The output stage as claimed in claim 1, wherein: the first current compensation circuit including a first current mirror coupled with the first current driving stage to compensate for the first bias current; and the second current compensation circuit including a second current mirror coupled with the second current driving stage to compensate for the first bias current.
 5. The output stage as claimed in claim 4, wherein: the first current mirror including at least a first current control transistor and a first current mirror output transistor each having a collector and base; the base of the first current control transistor couples with the collector of the first current control transistor and with the base of the first current mirror output transistor such that a collector current of the first current mirror output transistor provides the compensation for the first bias current; the second current mirror including at least a second current control transistor and a second current mirror output transistor each having a collector and base; and the base of the second current control transistor couples with the collector of the second current control transistor and with the base of the second current mirror output transistor such that a collector current of the second current mirror output transistor provides the compensation for the second bias current.
 6. The output stage as claimed in claim 1, where: the first current compensation circuit includes a sink transistor coupled with the first current driving stage to sink a first driving stage total collector current from the first current driving stage.
 7. The output stage as claimed in claim 1, where: the second current compensation circuit includes an source transistor coupled with the second current driving stage to supply a second driving stage total collector current from the second current driving stage.
 8. The output stage as claimed in claim 7, wherein: the source transistor is an NPN source transistor; the first current compensation circuit includes a PNP sink transistor coupled with the first current driving stage to sink a first driving stage total collector current from the first current driving stage; the PNP sink transistor configured to define a first compensation current to compensate for the first bias current; and the NPN source transistor configured to define a second compensation current to compensate for the second bias current.
 9. The output stage as claimed in claim 8, where: the first current compensation circuit including a first current mirror coupled with the PNP sink transistor and the first current driving stage, and configured to receive the first compensation current and to mirror the first compensation current at the first current driving stage; and the second current compensation circuit including a second current mirror coupled with the NPN source transistor and the second current driving stage, and configured to receive the second compensation current and to mirror the second compensation current at the second current driving stage.
 10. The output stage as claimed in claim 1, where: the output circuit includes at least one output source transistor coupled with at least one output sink transistor such that the output of the output stage is defined between the output source and sink transistors.
 11. The output stage as claimed in claim 1, where: the first bias current is about equal to a total collector current of the first current driving stage divided by a PNP beta factor.
 12. The output stage as claimed in claim 11, where: the second bias current is about equal to a total collector current of the second current driving stage divided by an NPN beta factor.
 13. The output stage as claimed in claim 1, further comprising: a clamping stage coupled with the input stage output and the output circuit, and configured to maintain the output level of the output stage to be approximately equal to the input stage output.
 14. The output stage as claimed in claim 13, where: the clamping stage including a clamping source transistor coupled with a clamping sink transistor, such that the input stage output is coupled to each of the clamping source and clamping sink transistors.
 15. The output stage as claimed in claim 1, further comprising: a feedforward path coupled between the input stage output and the output stage output, and configured to stabilize the output stage output at high frequencies.
 16. The output stage as claimed in claim 1, wherein: the first and second compensation circuits are configured to compensate for temperature effects of the output stage.
 17. The output stage as claimed in claim 1, wherein: the first and second compensation circuits are configured to compensate for process variations of the output stage.
 18. An amplifier output stage, comprising: a first current compensation circuit coupled with an amplifier input stage output and a first driving stage, and configured to compensate for a first bias current of the first driving stage at the amplifier input stage output to minimize reflection current back into the amplifier input stage; and a second current compensation circuit coupled with the amplifier input stage output and a second driving stage, and configured to compensate for a second bias current of the second driving stage at the amplifier input stage output to further minimize the reflection current back into the amplifier input stage.
 19. The amplifier output stage as claimed in claim 18, wherein: a first driving stage coupled with the amplifier input stage output and an output circuit of the amplifier output stage, the first driving stage configured to provide a first driving current to the output circuit; a second driving stage coupled with the amplifier input stage output and the output circuit of the amplifier output stage, the second driving stage configured to draw a second driving current from the output circuit.
 20. The amplifier output stage as claimed in claim 19, wherein: the output circuit of the amplifier output stage including: at least one source transistor coupled with the first driving stage, configured to receive the first driving current from the first driving stage and to source current to the output; and at least one sink transistor coupled with the second driving stage, configured to supply the second driving current to the second driving stage and to sink current from the output.
 21. The amplifier output stage as claimed in claim 18, wherein: the first driving stage including a first current multiplier coupled with the output circuit to provide a sufficient current to the output circuit while maintaining the first bias current at a substantially lower level reducing the amount of reflection current supplied back into the amplifier input stage; and the second driving stage including a second current multiplier coupled with the output circuit to draw a sufficient current from the output circuit while maintaining the second bias current at a substantially lower level reducing the amount of reflection current supplied back into the amplifier input stage.
 22. A method of enhancing an output gain, comprising the steps of: receiving an input stage output; and reducing an reflection current including: compensating for at least one bias current.
 23. The method as claimed in claim 22, wherein: the step of reducing a reflection current including: reducing a driving current needed to drive a stable output.
 24. The method as claimed in claim 23, wherein: the step of reducing the driving current needed including multiplying the driving current.
 25. The method as claimed in claim 22, wherein: the step of compensating for at least one bias current including generating at least one compensation current to compensate for the bias current.
 26. The method as claimed in claim 25, wherein: the step of generating at least one compensation current including: generating a first compensation current compensating for a PNP beta factor; and generating a second compensation current compensating for an NPN beta factor.
 27. The method as claimed in claim 25, wherein: the step of generating at least one compensation current including compensating for a mismatch between PNP and NPN beta factors.
 28. The method as claimed in claim 22, further comprising the step of: feedforwarding the input stage output to the stable output at frequencies above a predefined limit.
 29. The method as claimed in claim 22, further comprising the step of: clamping the stable output if the input stage output and the stable output diverge by a predefined limit.
 30. The method as claimed in claim 22, further comprising the step of: reducing a quiescent current maintaining the output stage in an active state without adversely affecting the output voltage.
 31. The method as claimed in claim 30, further comprising: the step of reducing the quiescent current including: reducing the amount of at least one bias current; and maintaining the output voltage by multiplying the driving current.
 32. The method as claimed in claim 22, wherein: the step of generating at least one compensation current including utilizing a current mirror circuit.
 33. An amplifier output stage configured to receive an amplifier input stage output and to generate a stable output to a variety of loads, the amplifier output stage comprising: a first and second current driving stage and an output circuit; the first driving stage coupled with the amplifier input stage output and the output circuit, and configured to generate a first driving current proportional with a first bias current based on the amplifier input stage output, to drive the output circuit; and the second driving stage coupled with the amplifier input stage output and the output circuit, and configured to generate a second driving current proportional with a second bias current based on the amplifier input stage output, to drive the output circuit.
 34. The amplifier output stage as claimed in claim 33, wherein: the first current driving stage includes a first current multiplier coupled with the output circuit, and configured to drive the output circuit at a predefined level while limiting the first bias current; and the second current driving stage including a second current multiplier coupled with the output circuit, and configured to drive the output circuit at a predefined level while limiting the second bias current.
 35. The amplifier output stage as claimed in claim 34, wherein: the first multiplier includes at least a first current multiplier transistor, the second current multiplier includes at least a second current multiplier transistor, each of the first and second multiplier transistors having a base, collector and emitter; the emitter of the first current multiplier transistor coupled with the output circuit to supply a first drive current to drive the output circuit; and the emitter of the second current multiplied transistor coupled with the output circuit to draw a second drive current to drive the output circuit.
 36. An amplifier output stage configured to receive an amplifier input stage output and to generate a stable output to a variety of loads, the amplifier output stage comprising: a first and second current compensation circuit coupled with an amplifier input stage output; the first current compensation circuit coupled with an amplifier input stage output, and configured to compensate for a first bias current to limit a reflection current back into the amplifier input stage; and the second current compensation circuit coupled with the amplifier input stage output, and configured to compensate for a second bias current to further limit the reflection current back into the amplifier input stage.
 37. The amplifier output stage as claimed in claim 36, wherein: the first current compensation circuit including a first current mirror to compensate for the first bias current; and the second current compensation circuit including a second current mirror to compensate for the first bias current.
 38. The amplifier output stage as claimed in claim 37, wherein: the first current mirror including at least a first current control transistor and a first current mirror output transistor each having a collector and base; the base of the first current control transistor couples with the collector of the first current control transistor and with the base of the first current mirror output transistor such that a collector current of the first current mirror output transistor provides the compensation for the first bias current; the second current mirror including at least a second current control transistor and a second current mirror output transistor each having a collector and base; and the base of the second current control transistor couples with the collector of the second current control transistor and with the base of the second current mirror output transistor such that a collector current of the second current mirror output transistor provides the compensation for the second bias current.
 39. The amplifier output stage as claimed in claim 36, where: the first current compensation circuit includes a PNP sink transistor to sink a first total collector current; and the second current compensation circuit includes an NPN source transistor to supply a second total collector current. 